1. Field of the Invention
The present invention relates to the field of switched mode power supplies, and more specifically to the field of d.c.-d.c.-type inductive storage voltage converters. Such converters, which enable to transform d.c. voltages, have many industrial applications, especially in the aeronautic field, where they are used to generate, from the d.c. electric network of aircrafts, voltages of 5V, .+-.12V, .+-.15V, . . . for supplying on-board electronic equipment.
2. Discussion of the Related Art
Generally, the operation of inductive storage converters is based on the implementation of energy transfer cycles including a period of accumulation of magnetic energy in an inductive component, via a primary circuit, followed by a period of restitution of this energy into a load to be supplied, via a secondary circuit. A "buck-boost" converter is a converter, the inductive component of which is a single-winding inductance, and a "fly-back" converter is a converter, the inductive component of which is a transformer including at least two windings.
In this field, the present invention more specifically relates to bidirectional voltage converters, which can transfer energy from the primary to the secondary as well as from the secondary to the primary. This type of bidirectional converter is particularly well suited to the supplying of complex loads (capacitive and/or inductive loads), accumulators or, further, reversible devices such as electric motors, which are likely to send energy back to the converter.
Examples of inductive storage bidirectional converters are described in U.S. Pat. No. 3,986,097, relating to converters of "fly-back" type, as well as in U.S. Pat. No. 4,736,151 and European patent application 336725, relating to converters of "buck-boost" type.
A general object of the present invention is to improve these bidirectional converters, the operation of which will first be described.
FIG. 1a shows the basic diagram of a conventional bidirectional "buck-boost" converter. In such a converter, a primary circuit and a secondary circuit are arranged on either side of an inductance L. The primary circuit includes a diode Dp and a chopping switch Tp assembled in parallel, the assembly being interposed between a supply voltage source Vin and inductance L. Similarly, the secondary circuit includes a diode/chopping switch assembly Ds/Ts interposed between inductance L and an output capacitor Cout. Capacitor Cout provides the smoothing of the output voltage Vout of the converter, applied to a load Z to be supplied. In practice, switches Tp, Ts, are electronic switches, such as MOS transistors or bipolar transistors.
As mentioned previously, the operation of the converter occurs in cycles of two periods, each cycle including a first so-called accumulation period, of duration Ton, when a current Ip flows through the primary, and a second so-called restitution period, of duration Toff, when a current Is flows through the secondary. FIG. 2 shows the total current I flowing through inductance L for the duration of a cycle. In bidirectional operation mode, so-called accumulation period Ton actually includes a first phase Ton1 of restitution in voltage source Vin of an excess energy accumulated in inductance L during a previous cycle, followed by a phase Ton2 of effective accumulation of magnetic energy in inductance L. FIGS. 1b and 1c show equivalent diagrams of the converter during phases Ton1, Ton2. During Ton1, inductance L generates current Ip which is negative (diode Dp conducts, FIG. 1b). When this energy has been entirely restituted, accumulation phase Ton2 begins, where Ip is positive (Tp is on, diode Dp is blocked, FIG. 1c). Similarly, so-called restitution phase Toff first includes a phase Toff1 of effective restitution in capacitor Cout and load Z of the energy stored by inductance L during Ton2, followed by a phase Toff2 of accumulation in inductance L of an excess energy supplied to capacitor Cout or to load Z during Toff1. FIGS. 1d and 1e show equivalent diagrams of the converter during phases Toff1, Toff2. It can be seen that secondary current Is is positive during Toff1 (diode Ds conducts, FIG. 1d), and then negative during Toff2 (Ts on, diode Ds blocked, FIG. 1e), load Z or capacitor Cout acting as a voltage generator. The energy stored during Toff2 is transferred to voltage source Vin during phase Ton1 of the following cycle, which characterizes a bidirectional operation.
A disadvantage of this type of converter is that its efficiency decreases as the operating frequency increases, whereas, contradictorily, it is preferable to choose high operating frequencies, from around 100 kHz to 1 MHz, to reduce the size and the bulk of the converter.
It is well known that the decrease in efficiency with the increase of the operating frequency is especially caused by energy losses in the switches during the switching periods. It should be reminded that the energy lost in a switch during a switching is equal to the product of the voltage across the switch by the current which runs therethrough and the switching time. The switching-on losses have to be distinguished from the switching-off losses. In a converter operating according to the bidirectional mode such as shown in FIG. 2, the problem of switching-on losses of switches Tp, Ts, is solved in principle since each switching on is preceded by a period of conduction of diode Dp, Ds, which guarantees a closing voltage close to 0 (diode voltage). Conversely, each opening of switch Tp, Ts, causes a reaction of the inductive component and an abrupt rise of the voltage across the switch, which increases the switching time of the switch by Miller effect. This phenomenon causes an energy loss in the switches which is all the more substantial as it happens many times in a second when the frequency is high. Moreover, the abrupt rise of the voltage upon switching off causes an emission of parasitic electromagnetic radiation. The same disadvantages also appear in a transformer converter of "fly-back" type, which operates according to the same principle.
To overcome this disadvantage, low loss converters have been provided, where the voltage edges appearing upon opening of the primary and secondary switches are smoothed due to the addition of so-called "smooth switching" capacitors.
FIG. 3 shows a low loss "fly-back"-type converter 10. Converter 10 is for example similar to the converter illustrated in FIG. 15 of European patent application 336725. It differs from the converter of FIG. 1 by the addition of two smooth switching capacitors Cp, Cs, and by a specific operating mode which involves two transition periods which will be described hereafter. Capacitor Cp is added in parallel to the diode/switch assembly Dp/Tp of the primary and capacitor Cs is added in parallel to the diode/switch assembly Ds/Ts of the secondary. Further, as the converter is of "fly-back" type, inductance L of FIG. 1 is replaced with a transformer 1 including a primary inductance Lp formed by a winding of Np turns of coil, and a secondary inductance Ls formed by a winding of Ns turns of coil.
The operation of the converter is illustrated by FIG. 4. FIGS. 4a and 4b show control signals Hp and Hs respectively applied to switches Tp and Ts. FIGS. 4c and 4d show the currents Ip and Is respectively flowing through primary inductance Lp and secondary inductance Ls of the transformer. FIGS. 4e and 4f show voltages VTp and VTs across switches Tp and Ts. FIGS. 4g and 4h show charge or discharge currents Icp and Ics of capacitors Cp and Cs. Finally, FIGS. 4i and 4j show parasitic currents Iop and Ios, which add to the currents of the primary and the secondary Ip and Is shown in FIGS. 4c and 4d.
As it can be seen in FIG. 4, each operating cycle of converter 10 includes four distinct periods which will be designated by their respective durations T1, T2, T3, T4.
Periods T1 and T3 are similar to periods Ton and Toff previously described, except for the fact that the respective currents of the primary Ip and of the secondary Is flow through distinct windings Lp and Ls. Thus, period T1 includes a restitution phase on the primary side where current Ip is negative (FIG. 4c, diode Dp conducting, Tp off or on), followed by an accumulation phase where current Ip is positive (diode Dp blocked, Tp on, Hp=1). Conversely, period T3 first includes a restitution phase on the secondary side where current Is is positive (diode Ds conducting, Ts off or on) followed by an accumulation phase where current Is is negative (FIG. 4d, diode Ds blocked, Ts on, Hs=1). As shown by dotted lines on FIGS. 4a, 4b, during periods T1 and T3, there is a margin of liberty for closing switches Tp and Ts, as long as diodes Dp and Ds are conducting.
Periods T2 and T4 are short duration transition periods during which Tp and Ts are maintained off. In FIG. 4, T2 and T4 are not shown to scale, and are in fact of the order of one tenth or one hundredth of T1 and T3. During these transition periods, capacitor Cp discharges and capacitor Cs charges, and conversely (FIGS. 4g and 4h). Diodes Dp, Ds, are in the blocked state, the energy stored in the transformer is neither transferred to the primary, nor to the secondary.
Those skilled in the art will note that the smooth switching capacitors Cp, Cs, should not be confused with the capacitors that are found is so-called resonance converters, the operating principle of which is not comparable with that of inductive storage capacitors, which are the subject of this description. Here, capacitors Cp, Cs, are not means enabling energy transfer through the converter. They are, conversely, chosen to have small charge or discharge times T2 and T4 with respect to times T1 and T3 during which the energy transfers are performed in the converter.
The advantage of adding transition periods T2, T4, and of involving smooth switching capacitors Cp, Cs, is that upon opening of a switch Tp or Ts, the capacitor Cp or Cs associated with the switch charges progressively and prevents the abrupt rise of voltage VTp or VTs. As shown in FIGS. 4e and 4f, voltage VTp or VTs of the switch will increase during transition period T2 or T4 until it reaches its maximum value, equal to Vin+Vout*Np/Ns for VTp, and Vout+Vin*Ns/Np for VTs. The switching-off losses due to the Miller effect are thus suppressed, or at least considerably reduced.
These converters however have other disadvantages, which will be now described.
Disadvantages of the use of smooth switching capacitors
The applicant has first noticed that despite the smoothing of the rising/falling edges of VTp and VTs, the switchings of the switches remain a source of parasitic electromagnetic radiation. More specifically, it has been noted that the radiation is generated by parasitic currents Iop, Ios, due to a parasitic oscillation phenomenon between smooth switching capacitors Cp, Cs, and parasitic inductances present in the converter, which can be represented as a first inductance lp in series with Lp and a second inductance ls in series with Ls (FIG. 3). Currents Iop and Ios, shown in FIGS. 4i and 4j, appear after each switching of switches Tp, Ts, follow an oscillating state which damps slowly, and exhibit a peak intensity which can be equal to the maximum values reached by Ip and Is at the end of T1 and T3. Current Iop flows through the entire loop formed by the primary circuit, including inductance Lp, the diode/switch/capacitor assembly Dp/Tp/Cp and voltage source Vin, as well as the electric connections between these components. Similarly, current Ios flows through the entire loop formed by the secondary. Thus, each primary or secondary loop emits electromagnetic radiation due to an antenna effect, proportional to a magnetic flow .PHI. expressed as .PHI.=S+B, with B representing the magnetic field created by the parasitic current and S being the surface of the loop.
A first object of the present invention is to attenuate the effect of such parasitic currents, and to provide a converter structure with low radiation.
To achieve this object, the present invention provides to modify the arrangement of the smooth switching capacitors, and to connect them in parallel across the inductive component of the converter. The advantage is that, on the one hand, the smooth switching function of the capacitors is kept and, on the other hand, the smooth switching capacitors form, with the primary and secondary inductances, loops with reduced surfaces and lengths wherein are confined the two oscillation currents Ios, Iop. Thus, magnetic flow .PHI. and the parasitic radiation are considerably reduced.
Disadvantages of conventional control systems
Generally, the voltages issued by bidirectional converters are stabilized via a control system of the primary and secondary switches Tp and Ts, which permanently monitors output voltage Vout and compares it with a reference voltage Vref. The desired aim is to maintain output voltage Vout constant, by controlling switches Tp and Ts and durations T1 and T3 of conduction in the primary and the secondary.
In a converter in stabilized state, times T1 and T3 are linked by a general relation which can be found as follows:
(a) during T1, the height .DELTA.Ip=Ip2-Ip1 of the current ramp which flows through the primary winding Lp is expressed as: EQU .DELTA.Ip=Vin*T1/Lp (1) PA1 where Ip2 is the positive current at the end of period T1, Ip1 the negative current at the beginning of period T1 (FIG. 4c), Vin is the input voltage, and Lp the inductance of the primary winding. PA1 (b) in an equivalent way, during T3, the height .DELTA.Is=Is1-Is2 (FIG. 4d) of the current ramp flowing through the secondary winding Ls is expressed as: EQU .DELTA.Is=Vout*T3/Ls (2) PA1 (c) the general principle of continuity of energy E in a transformer enables to write that the stored energy at the end of a phase is equal to the transferred energy at the beginning of the next phase. This leads to the following relations: E=1/2*Lp*(Ip2).sup.2 =1/2*Ls*(Is1).sup.2 for the transition from T1 to T3, and E=1/2*Ls*(Is2).sup.2 =1/2*Lp*(Ip1).sup.2 for the transition from T3 to T1. Since, besides, in a transformer, Lp=Al*Np.sup.2 and Ls=Al*Ns.sup.2, Al being a constant, it can be inferred therefrom that: EQU Np*.DELTA.Ip=Ns*.DELTA.Is, (4) PA1 (d) by combining this latter relation with expressions (1) and (2) of .DELTA.Ip and .DELTA.Is given at (a), it can be inferred that: ##EQU1## PA1 an optimal efficiency on a wide range of values of input voltage Vin, in the case of a voltage Vin which is not constant, PA1 the possibility of setting output voltage Vout on a wide range of values while maintaining an optimal efficiency, PA1 the monitoring of a single side of the circuit, to limit the number of current sensors. This will be preferably current Ip on the primary side, for practical reasons: the detection of a possible overcharge problem and the control of the starting upon power-on, PA1 a totally symmetrical operation: the possibility of accepting at the output a load issuing current.
This relation gives the ratio which exists between input voltage Vin and output voltage Vout, this ratio being assumed to be constant in the case of a converter in stabilized state. Of course, this relation concerns a "fly-back" transformer converter: in the case of a "buck-boost" converter, Ns/Np=1.
Relation (5) can also be written as follows: ##EQU2##
Thus, the ratio T3/T1 is always constant for a stabilized output voltage Vout and a constant input voltage Vin, whatever the control method used.
Examples of control systems based on this general operating principle are described in above-mentioned European patent application 336725, in relation with FIGS. 12, 14, 15 of this application. According to prior art, it is known to control the duration of the closing of primary switch Tp based on an error signal .epsilon. generated by comparing output voltage Vout with a reference voltage Vref. It is also known to trigger the opening of secondary switch Ts (which marks the end of period T3) when current Is flowing through the secondary exceeds a given threshold (current control at the secondary) or when output voltage Vout falls beyond a given value.
One of the disadvantages of known control systems is that they do not give an optimal efficiency on a wide range of values of input voltage Vin. FIG. 5 illustrates the curve of efficiency according to the variations of the input voltage of a converter controlled in a conventional way. It can be seen that the efficiency, which is optimal for a nominal value of the input voltage, deteriorates as Vin increases. This phenomenon is attributed to an increase in the amplitude of current ramps .DELTA.Ip, .DELTA.Is, in the converter and to losses by Joule effect. This disadvantage becomes particularly disturbing when there is no stable supply source. For example, in the case of an aircraft, the nominal voltage Vin distributed by the aircraft network is around 28V but can vary between 12V and 36V or more (up to 80V in the case of a network failure). In such operating conditions, the efficiency of a converter is difficult to control.
Another disadvantage of some known control systems is that they are based on both a monitoring of the primary current Ip and a monitoring of the secondary current Is, and that they require at least two current sensors, one being placed at the primary, the other at the secondary.
A second object of the present invention is to provide a control system of a bidirectional converter which offers the following advantages:
Disadvantages of the current sensors used for monitoring the currents flowing through a voltage converter
As indicated previously, the control of a converter requires a monitoring of the currents flowing through the converter. This monitoring can be performed by current sensors. Given the high operating frequencies and the possibly high currents, low loss current sensors with a short response time are desired.
FIG. 6 shows a current sensor 20 conventionally used in voltage converters or other similar devices. This voltage sensor includes a transformer 21 which has a primary winding Wp, run through by a current ip to be measured, and a secondary winding Ws for measuring this current. The secondary winding Ws is connected, via a diode 22, to a measuring resistor r. Finally, an impedance 23 for demagnetizing transformer 21 (for example a Zener diode or a resistor with a high value) is connected in parallel across Ws. During a measurement period, current ip flows through winding Wp and a current is proportional to ip appears in winding Ws, the ratio between ip and is being determined by the ratio ns/np between the number of turns of coil of Ws and Wp. Current is flows through diode 22 and creates, across measuring resistor r, a voltage V=r*is representative of the current ip to be measured. This measurement period must necessarily be followed by a rest period, where current ip has to be zero, so that transformer 21 demagnetizes. During the rest period, the demagnetization is performed by impedance 23, across which a voltage with a reverse biasing appears. Without the rest period, an increasing error voltage would appear at the output of the current sensor, due to the storage of a magnetizing current in transformer 21 and to the saturation thereof.
The main disadvantage of this current sensor is that it can only detect a current in a single direction, imposed by diode 22. Besides, the magnetic core of the sensor very rapidly reaches saturation when a reverse current with respect to the normal detection direction flows through primary winding Wp. This is particularly disturbing when the positive values of primary current Ip are desired to be measured in a bidirectional converter. Since current Ip is negative before being positive, the sensor is first run through by a reverse current with respect to the normal direction of detection, which rapidly leads the magnetic core of transformer 21 to saturation. When, afterwards, current Ip becomes positive, the output voltage of the sensor, instead of being proportional to Ip as it could be expected, is in fact tainted with error.